Eric Bogatin's Blog: What I Learned This Month

Next No Myths Allowed Webinar, “Selecting Capacitor Values for the PDN”, Dec 9  2009, 1 pm EDT. Signup now.

“There ain’t no such thing as a free lunch,” is how the phase, popularized in Robert Heinlein’s book, The Moon as a Harsh Mistress, usually goes.

But that’s not how Scott McMorrow, director of engineering at Teraspeed, uses the phrase. He is fond of saying “There ain’t no such thing as a free launch.”

It’s sort of ironic, because he and his team are world experts at providing nearly free launches.

A launch is a transition from one transmission line geometry to another. While a coax cable and a stripline in a circuit board may each be electrically transparent, when one transitions into the other, the interface, or launch, will always show up as a discontinuity.

This discontinuity will cause a reflected signal and a reduction in the transmitted signal, which shows up in the insertion loss. The larger the discontinuity, the bigger the impact on the insertion loss. And, due to the physical size of a launch, it is always more of a problem at higher frequencies.

To minimize the launch discontinuity, Teraspeed recommends using a surface mount SMA connector and carefully optimizing the features of the launch via pad stack.

The figure to the left shows the TDR response of a conventional, well designed launch and a Teraspeed “free launch”, with a roughly 35 psec rise time signal and 5 Ohms per division. This was reported, most recently, at DesignCon 2009 and can be found in the reprinted article on the Simberian web site.

The pad stack includes the capture pads, the via barrel diameter, location of return vias, and clearance holes in any planes. Of course, what works in one board will be not always be the best design in another board due to the precise combination of signal layer and plane layer assignments and dielectric thicknesses.

Translating a specific board’s pad stack into the virtual world of a 3D field solver enables you to quickly optimize the clearance holes for a transparent launch. For example, if the launch impedance is high, make the clearance holes smaller. If the impedance is low, make the clearance holes larger.

This principle of a “free launch” applies to all transitions, especially important from the planar geometry of a circuit board to the 3D geometry of a connector.

Samtec made popular the term, “the final inch” to describe the break out region (BOR) of a circuit board connector’s via field. Using this principle of optimizing a few features in the immediate region of the launch, they can make the circuit board transitions into their connectors nearly transparent.

When done well, the transition from any connector to board traces can be transparent. This is important when designing test vehicles, ATE load boards and high performance product boards. As PCIe and USB enter the 5 Gbps and above regime, designing transparent launches will be an important skill.

For information on this and other multi gigabit topics, check out our new class, Multi Giga Bit Design (MGBD).

Hope to see you in cyber space at our next webinar!

Next No Myths Allowed Webinar, “Selecting Capacitor Values for the PDN”, Dec 9  2009, 1 pm EDT. Signup now.

Our last webinar, “Stack up Design of Differential Pairs”, had over 600 registered viewers. If you missed it, you can still watch the recording for free.  The positive response was overwhelming. Surprisingly, most of the emails I got back were about a comment I made in passing.

“The single-ended impedance of one line does not change as an adjacent, grounded line is brought closer.” This is in contrast to the odd mode impedance of one line which will decrease as an adjacent line is brought closer.

Many people said they did not think the single-ended impedance would really be constant- it should decrease. Colin Warwick and Steve Martin, independently, went to the effort of running a simulation to check this out.

Steve did a super analysis and sent me the plot to the left which shows what is going on. In his example of two coupled microstrips, the line widths were 300 microns, which is 12 mils. The dielectric thickness he used was 254 microns, or 10 mils. This comes out as a relatively high impedance line, about 70 Ohms, single-ended.

He calculated the odd mode impedance, the even mode impedance and the single-ended impedance of one line as the other line was brought closer.

The center trace in the plot, the green line, is the single ended impedance. It is absolutely constant until a spacing of about 300 mils, which is the line width. Closer than this tight a coupling, the single-ended impedance begins to drop. But, as long as the spacing is greater than the line width, what we would call tightly coupled, the single-ended impedance is constant.

It is only when the two lines are much closer together than tightly coupled, does the single-ended impedance drop. Up until this extreme tight coupling, the single-ended impedance of a line really is constant.

Hope to see you in cyber space at our next webinar!

Next public classes: Essential Principles of Signal Integrity, Advanced Signal Integrity Design, and Multi GigaBit Design, Sept 29- Oct 7 in San Jose, CA.

Next No Myths Allowed Webinar, “Stack-up Design for Differential Pairs”, presented free on Sept  16, 1 pm EDT.

In all the lectures I give on differential pairs and differential pair design, I always say that you should forget the words “differential mode” or “common mode.” Using these terms confuses the use of the words and, I think, significantly confuse our intuition and make it even harder to understand this already confusing topic.

The words we use influence the way we think and the models we carry around inside. Instead of differential mode, we should be using the words differential signals and common signals, or odd mode and even mode.

If you are consistent  using these terms, you will keep clear the concepts of the differential signal component and the common signal component, as properties of the signals, and the odd and even modes as properties of the interconnects.

An EMC engineer buddy of mine told me my quest to change the words we use in the industry is like Don Quixote, tilting at windmills, which Wikipedia defines as “fighting unwinnable or futile battles”.

OK, so I have given up on my unwinnable quest to change the industry, but I still think if you want to calibrate your intuition and really understand what goes on when a differential or common signal propagates down a differential pair and it sees a differential impedance or a common impedance, don’t use the words differential mode or common  mode. It is not needed and will screw up your intuition.

If you want to really understand differential pairs and how to design them, join me for our next No Myths Allowed webinar, Sept 16, 2009. Sign up on www.bethesignal.com

Hope to see you in cyberspace!

Next public classes: Essential Principles of Signal Integrity and Advanced Signal Integrity Design, and Multi GigaBit Design, Sept 29- Oct 7 in San Jose, CA.

Next No Myths Allowed Webinar, “Stack-up Design for Differential Pairs”, presented free on Sept  16, 1 pm EDT.

I was not able to attend the 2009 IEEE EMC Symposium held in Austin last week, but as a member of the EMC Society, I got a copy of the proceedings.  I was delighted to find a number of really exciting papers. Some introduced new ways of modeling or evaluating interconnect structures like vias and the power-ground cavity, while others recounted simple experiments to verify or quantify well know solutions.

Over the next few months, I will try to review some of the more relevant paper to give them a little more visibility.

Two papers in particular showed very simply the problem with connectors in cable assemblies. In the ASID class, I spend a whole module on EMI problems and solutions. While external cables, even shielded coax cables, may act as the radiating antenna, its not the cable that is the source of the noise voltage that drives the common currents that radiate; it’s the connector.

In the paper, “Relationship Between Connector Contact Points and Common-mode Current on a Coaxial Transmission Line,” by Hayahi, Mizuki and Sone, of the Tohoko University in Sendai, Japan, the authors illustrate this principle with a simple construction.

They built a two-section coax cable, with the sections connected in a region where they could vary the number of connecting wires between the shields. Four configurations were constructed, with 1, 2, 3 and 4 connecting wires between the shields.

What I teach in the ASID class is that the more the connector allows the return current to flow symmetrically around the signal current, the more cancellation of external magnetic field lines and the lower the total inductance of the return path. In these four different return path configurations, the more the connection looked like a 360 degree, symmetric path, the lower the common currents around the cable and the lower the emissions.

Though this was not a profound conclusion, nor unexpected, it was a simple, beautiful example of this principle that the connector is often the weak link in radiated emissions from coax cables.

In the paper, “Effectiveness of Shield Termination Techniques Tested with a TEM Cell and Bulk Current Injection,” by Bradley and Hare of NASA Langley Research Center in Hampton, VA, the authors show by direct measurement the radiated emissions from cable assemblies with different shield termination schemes.

When a pigtail or drain wire is used to provide the return path connection, the radiated emissions are 20 dB higher than with a connection that goes 360 degrees around the signal current. In fact, the authors compare four different ways of making the 360 degree termination, using foil, clamping the braid, soldering the braid and an elaborate clamp to the backshell with additional braid overlapping the cable shield.

They find that all the methods that provide 360 degree connection work equally as well.

These two papers did not rock the world, but they demonstrate in simple, clear experiments this principle that when the return currents are not symmetrical around the signal path, the noise across the total inductance of the return path will drive common currents which will radiate.

If you want to learn more about solving EMI problems by understanding the root cause of the problem and the essential principles on which the solutions are based, check out the classes and lectures listed on our web site.

Check out our next public classes: Essential Principles of Signal Integrity and Advanced Signal Integrity Design, Oct 11-14 in Hillsboro, OR.

Check out our next No Myths Allowed Webinar, “Stack-up Design for Differential Pairs”, presented free on Sept  16, 1 pm EDT.

The design process is a creative process and intuition is the most important skill you rely on first. Once you have a design, then you apply your analysis skills to evaluate the cost-performance tradeoffs.

Wikipedia defines intuition as the “ability to sense or know immediately without reasoning”. I like to think of intuition as what you take away after you have done the reasoning- what has become ingrained in your understanding and you trust to use every day. It becomes your “common sense” of the right way of doing things.

The better your intuition or common sense about signal integrity, the better your first pass design will be. This translates to a shorter and lower cost design process.

Because signal integrity is sometimes “anti-intuitive”, we sometimes have to “re-calibrate” our intuition for these analog electromagnetic effects important in interconnects. The Essential Principles class I teach, is really about building a strong foundation for your common sense about signal integrity. Here in an abbreviated list, are 13 of the most important principles to strengthen your common sense when dealing with signal integrity:

1.    The fastest way of fixing a problem is by first identifying its root cause and then applying the Youngman Principle.

2.    All interconnects are transmission lines, no matter how long or how short they are.

3.    All signals are dynamic and constantly move along the transmission line at the speed of light in the dielectric, roughly 6 inches/nsec.

4.    Signals sees an instantaneous impedance each step they takes along a transmission line.

5.    The return current is exactly coincident with the signal current, flowing in the opposite direction, in the return conductor.

6.    Reflections occur whenever the instantaneous impedance changes.

7.    Don’t confuse the distributed cross talk between transmission lines which rarely extends beyond adjacent lines, with ground bounce cross talk which can extend to many adjacent transmission lines.

8.    Ground bounce occurs due to the dI/dt of the return current passing through the total inductance of the return path.

9.    The differential impedance of a differential pair can be just as well controlled for a tightly coupled as loosely coupled pair.

10.    A real capacitor behaves like a series combination of ideal RLC elements even up to the GHz bandwidth.

11.    Always try to place power and ground planes on adjacent layers with thin dielectric between them.

12.    Use SPICE to simulate the parallel resonances when multiple capacitors and the power and ground planes are connected in parallel.

13.    Assign return path layers and signal routing in the stack up based on the ability to provide a return via whenever a signal via changes layers.

If you want to learn more about common sense signal integrity, take our class, Essential Principles of Signal Integrity, attend one of our webinars or visit our web site, www.beTheSignal.com.

Check out our next public classes: Essential Principles of Signal Integrity and Advanced Signal Integrity Design, Oct 11-14 in Hillsboro, OR.

Check out our next No Myths Allowed Webinar, “Link Analysis with Return Path Discontinuities”, presented free on July 7, 1 pm EDT.

The pop quiz this month was, “When two adjacent signal lines transition from one signal layer, through a pair of planes, to another signal layer, the return current flows between the cavity formed by the planes. The impact of the return path discontinuity is strongest on which S-parameter term.”

At 28%, the consensus was S11, followed by SDD11 at 23%. This is surprising, as neither answer is correct. This topic is covered in detail in the SI-Insight for June, 2009, released this month, “Ground bounce in Vias.” We also touch on this topic in this month’s webinar, NMA-820, “Link Analysis with Return Path Discontinuities,” which, if you missed the free live event on July 7, can be viewed from the recording on the web site.

Which quality of the pair of lines is most affected by the return path discontinuity? The impedance discontinuity, which S11 is most sensitive to, is minor. And, as we show in the paper and the webinar, the differential signal is mostly immune to this return path discontinuity.

In fact, the biggest impact is on the single ended cross talk between the two lines. The correct answer to this month’s pop quiz is S31, the near end cross talk. Especially when the adjacent lines are far apart, the edge coupled cross talk can be very small, but the cross talk between the signal vias that pass through the cavity can be very large and long range.

The figure to the left, Figure 19 taken from the SI-Insights report, shows the measured near end cross talk between two signal lines, far apart for three design cases. The red measured response is with no signal vias, just two 50 Ohm lines about 15 lines widths apart. The green response is the measured near end crosstalk between these two lines routed from the top layer to the bottom layer, going through vias, with adjacent return vias.

The blue trace is the measured response of similar via transitions, but without the return vias. The coupling between the signal lines from noise injected in the cavity is more than 20 dB higher than with either no vias or vias with adjacent return vias.

Feel up for the challenge of a new pop quiz? Check out the new pop quiz on the web site.

Our next No Myths Allowed webinar, July 7, 1 pm EDT, “Link Analysis with Return Path Discontinuities”. Details and registration available at www.beTheSignal.com

“The space shuttle solid rocket boosters were designed the way they were because of a horse’s ass,” is the way the legend goes. There is actually some basis of fact in this legend. For the details, check out the article in snopes.com.

Even if only partially true, there is insight in this story that can be applied to many other cases of the origin of specs.

The story goes that the width of Roman chariots was defined in terms of the width of two horses side by side- basically the width of their asses. As the Romans conquered most of Europe and England, they brought their chariots, and their roads with them.

Over time, the wheels dug ruts. As new wagons were introduced, they were built with the same axle spacing because this is what the infrastructure supported and to be backward compatible with the existing ruts. After all, there is still a fundamental limit to how close you can get a couple of horses, so this axle size is not so unreasonable.

When the first railroads were built, the cars used were based on current carriage technology, which had an axle spacing of 4 ft, 8.5 inches. The early trains were even called “iron horses.” Once established, this standard proliferated and the railroad car axle and railroad tracks were standardized at 4 ft, 8.5 inches.

When Thiokol was designing the diameter of the solid fuel booster, so the legend goes, they had to limit its OD based on fitting on a railroad car, which is limited by the axle pitch, which is limited by a pair of horses asses.

Is 4 ft, 8.5 inches the ideal axle span spec? I have no idea. This spec is used to be backward compatible with the existing infrastructure. Unless you want to start from scratch, all rail car axles need to be this length. If there is no compelling reason otherwise, this spec may be acceptable.

When considering a new product, you should always ask two important questions: is it important to be backward compatible, and is there a compelling reason to consider alternative values, as in a better cost-performance balance?

There are two specs in the circuit board industry where it is important to consider these questions; board thickness and differential impedance.

I recently read a really interesting piece Lee Ritchey wrote in Circuitree Magazine about the origin of the board thickness spec of 063 mils.

He points out that the original “circuit boards” were made of plywood and were designed to provide a mechanical support for vacuum tubes and other large components. As the leads were short, the thinnest plywood was used, 1/16th of an inch, which is 063 mils. Plywood evolved into Bakelite, which evolved into fiber glass. The same thickness of 063 mils was used in each generation because that was what was used before. As edge card connectors were introduced, they were design for 063 which locked the standard in for compatibility.

With the introduction of many layers, they all would not fit in 063 mils so a new standard was introduced at 1.5 x 063 or 93 mils.

He rightly asks, is there a performance reason for 063? If you don’t need the thickness for mechanical support or edge connector compatibility, is it still the right thickness to use in your application? Not if it costs more.

The same question should be asked about 50 ohms single-ended or 100 Ohm differential impedance. As I wrote about recently, 50 Ohms had its origin 70 years ago as a way of minimizing attenuation in coax cables for radar and radio applications.

If your application is high speed serial links, you still have a need for optimized design for low loss, but in PCB geometries, 100 Ohms is not the optimum value. As we describe in some of our classes, lower impedance has lower loss. Intel has recommended 85 Ohm differential impedance for PCI gen II operating at 5 Gbps. It has lower loss and is a better match to the typically 70 Ohm differential impedance of through vias in a thick circuit board.

Many connector companies, such as Molex in their Impact series, are offering connector options at 85 Ohms. It will become the new standard.

This may not be the right answer for every design, as the lower differential impedance may mean higher power consumption. All of life is a trade-off and that’s why engineers need to be empowered to make their own decisions about their specific, custom products.

Sometimes it is useful to take a step back and look at why the specs we use are the value they are, and if there is a compelling reason, change them to help find a better cost-performance balance.

Our next No Myths Allowed webinar, July 7, 1 pm EDT, “Link Analysis with Return Path Discontinuities”. Details and registration available at www.beTheSignal.com

On Tues, I walked the floor of the Sensors Expo in Chicago and came away with a new understanding of the coming importance of sensor “motes”.

Motes, small, remote sensor nodes as part of a larger network of distributed sensors, have been around for a while. Pictured at left is an example of a mote from Powercast.

At this conference, I saw a harmonic convergence of four technologies and two important killer apps that I think will accelerate the development and implementation of sensor motes.

Energy harvesting or scavenging, is the technology of taking “waste” or local fluctuations in energy from the environment and collecting and storing it in a battery or large capacitor for later use.

I saw a number of companies showing off savaging techniques, leveraging, random vibrations, using piezo-electric transducers which convert vibration into voltage, tiny wind turbines which convert a gentle breeze into electricity, thermopiles which convert a small temperature gradient into a voltage, small solar cells which convert low light levels into voltage and even rectifiers that harvest energy from the local EM noise.

The local storage is a small, typically either thin film or polymer based solid state rechargeable battery that could be recharged thousands of times. Regulating the energy conversion and the charging as well as the output voltage is a tiny, ultra low power ASIC chip.

This combination acted as a local power source for an ultra low power microcontroller, with power consumption measured in the microwatt range. TI was showing off their MSP430 series microcontroller with less than 1 microA standby current.

The microcontroller measures the output from a variety of sensors such as humidity, temperature, voltage, vibration, light level, proximity, rotation, tilt, B field, or acceleration.

Connected to the microcontroller and using the local power source is a micro power wireless transceiver, using either a Zigbee, 802.11x or even a proprietary standard.

This combination of four technologies enables a standalone, remote sensor node which, with the right software, can self-assemble into a highly linked network communicating between each other and a base station over long distances. Once set up, the node never needs servicing, never needs a battery changed, never needs replacement. They can be located in remote, inaccessible locations and forgotten.

The two applications that were talked about the most at the conference were environmental monitoring in commercial buildings and monitoring of the future smart power grid. Both applications apply to improving our energy efficiency. Given the number of nodes that might be used in a building and the number of buildings, and the number of nodes along the proposed smart grid, the unit volume of motes could be in the billions.

Will this technology be what fuels the next killer app for the electronics industry?

Mark your calendars for our next No Myths Allowed webinar, scheduled for Wed, May 6, 2009 at 1 pm EDT. Like all of our webinars in this series, it will last about 45 minutes with 15 minutes for Q&A and be well worth your time.

This one is entitled, “S-Parameters, Signal Integrity and You.” In 45 minutes, we will introduce the most important features of S-parameters, starting at the very beginning and exploring some of their features and why they are becoming the defacto standard to describe the high frequency behavior of interconnects.

I’ve started a list of the questions we will address in our brief 45 minutes. If you have another question important to you not on the list, drop me a note and I will consider adding it to the webinar. Here is what we will cover:

• What are S-parameters?
• Why should I care?
• Where do they come from?
• How are they simulated?
• How are they measured?
• How accurate are they?
• How do I look at them?
• What’s the deal with differential S-parameters?
• How do I measure differential S-parameters?
• What can I learn about an interconnect from them?
• What can I do with them?
• What are some of the common pitfalls I should watch out for?
• What are some of the resources I can leverage to get more value from S-parameters?

You must sign up by May 4th, in order to attend. We will send out an email note with the access information the day before the webinar. The webinar will be recorded and posted to our website, along with all our other video recordings, and available to all paid subscribers the day after the presentation.

Hope to see you in cyberspace!

Our next No Myths Allowed Webinar: “S-Parameters, Signal Integrity and You”, May 6, 2009 1 pm EDT. Details are at www.beTheSignal.com

Last month’s pop quiz was:

“To build a transparent differential via, what is the most important feature to engineer?”

Here are the results from 180 participants.  While the most common answer to all signal integrity questions is “it depends”, it’s not always the best answer. In the case of transparent vias, the limitation is really the via stub. The correct answer is the fourth one, minimize the stub length, which 23% of you correctly answered.

All of the other factors are important. but the one with the biggest impact and which limits the bit rate of signals transmitted down the interconnect, is the length of the via stub. As a rough rule of thumb, the maximum stub length, in mils, that can be used in an interconnect system to transmit a bit rate, BR, in Gbps, is roughly:

Len < 300 mils/BR.

For the details on this and other properties of differential pairs, check out the last No Myths Allowed webinar, “NMA-800 Practical Differential Pair Design”. The handouts are available for download, and if you missed the live webinar, you can view the recording from our web site, www.beTheSignal.com.

And don’t miss our next No Myths Allowed webinar, “S-Parameters, Signal Integrity and You” on May 6 at 1 pm PDT.  I’ll see you there!